1. Field of the Invention
The present invention relates to an automatic gain control detection circuit.
2. Description of Prior Art
There is a conventional circuit for detecting an automatic gain control (AGC) signal, for example, a circuit as shown in FIG. 1. The conventional AGC detection circuit is constituted in a well-known double balanced type multiplier circuit. The double balanced type multiplier circuit is comprised of first, second and third differential amplifiers D1 to D3. Second and third differential amplifiers D2 and D3 are respectively connected in series with two current paths of first differential amplifier D1. Transistors Q1 and Q2 constituting the first differential amplifier D1 are respectively coupled at their bases to input terminals 11A and 11B. Transistors Q3 and Q4 constituting second differential amplifier D2 are respectively coupled at their collectors to output terminals 12A and 12B. Also, transistors Q5 and Q6 constituting third differential amplifier D3 are respectively coupled at their collectors to output terminals 12A and 12B.
A first reference voltage V1 is applied from a first reference voltage source E1 to the bases of transistors Q4 and Q5 through a resistor R3. A second reference voltage V2 is applied from a second reference voltage source E2 to the bases of transistors Q3 and Q6 through a resistor R4. And a power supply voltage V3 is applied from a power supply source E3 to the collectors of transistors Q3 and Q5 through a resistor R1 and the collectors of transistors Q4 and Q6 through a resistor R2. The base of transistor Q1 is connected to the bases of transistors Q3 and Q6 through a capacitor C1, while the base of transistor Q1 is connected to the bases of transistors Q4 and Q5 through a capacitor C2. An AGC detection output Vout is obtained between the common connection node of the collectors of transistors Q3 and Q5 and the common connection node of the collectors of transistors Q4 and Q6, i.e., between output terminals 12A and 12B. The common connection node of the emitters of transistors Q1 and Q2 is connected to a current source 13.
When an input signal voltage from an input signal source 11 is given as Vin and a current generated from current source 13 is given as Iee, collector currents Ic1 to Ic6 of transistor Q1 to Q6 in above circuit are expressed as follows. (Vt is a thermal voltage of the transistors.) ##EQU1## From Equations (1) to (6) ##EQU2##
Assuming a load resistance of a load circuit 12 to be connected between output terminals 12A and 12B as R1, AGC output Vout is expressed as follows. ##EQU3##
Where, Iout is an output current flowing through load circuit 12. If assuming Vin&lt;&lt;Vt, EQU Vout=R1.multidot.Iee.multidot.(Vin/2Vt).sup.2 ( 12)
When replacing input Vin to Vo sin t, ##EQU4##
When AGC output Vout is taken out via a low-pass filter (not shown), the following Equation is obtained. ##EQU5##
As described above, it will be understood that the AGC detection circuit can obtain AGC output Vout which is proportional to the square of DC voltage component Vo of input signal Vin and simultaneously inversely proportional to the square of thermal voltage Vt of the transistors.
The conventional AGC detection circuit described above obtains ABC output Vout in proportion to 1/Vt.sup.2. The thermal voltage Vt of the transistors can be expressed as follows. EQU Vt=k.multidot.T/q
where, k is the Boltzmann's constant, q is the element charge, and T is absolute temperature.
Therefore, the AGC output Vout is inversely proportional to the square of the absolute temperature T so that it varies excessively in accordance with temperature variation. Further, the DC potential of AGC output Vout becomes large so that it easily saturates to its maximum amplitude.
For solving such a problem, many improvements have been made for compensating above output variation by making the current Iee of current source 13 become proportional to the absolute temperature T.
As an example of the current source whose output current is proportional to the absolute temperature T, there is a circuit as shown in FIG. 2.
In FIG. 2, transistors Q11 and Q13 are same size and each emitter is grounded through resistors R11 and R13 respectively. Transistor Q12 has an emitter junction area that is N times larger than that of transistors Q11 and Q13. The emitter of transistor Q12 is connected to the emitter of transistor Q11 through a resistor R12. Transistors Q14, Q15, and Q16 and resistors R14 and R15 constitute a current mirror circuit and flow an equal current Io to the collectors of transistors Q11 and Q12. Also, transistors Q17, Q18, and Q19 and resistors R16 to R20 constitute a starter circuit.
Now, assuming Vbe11, Vbe12 and Vbe13 as voltages across the base and the emitter of respective transistors Q11, Q12 and Q13 and Is as a reverse direction saturation current of transistors Q11 and Q13, they have relations with each other as expressed below. EQU Vbe11=Vbe12+IoR12 (15) EQU Vbe11=Vbe13=Vt.multidot.ln (Io/Is) (16) EQU Vbe12=Vt.multidot.ln [Io/(NIs)] (17)
From Equations (15) to (17), ##EQU6##
Since Vbe11 and Vbe--are equal (see Equation (16)), the potential drops in resistors R11 and R13 are also equal. Also, since .beta.&gt;&gt;0, output current Iee(t) of the circuit, which flows through transistor Q13, is given from following Equation. EQU R13.multidot.Iee(t).congruent.R11.multidot.(2.multidot.Io)
Therefore, ##EQU7##
When substituting Vt=k.multidot.T/q, ##EQU8##
Accordingly, output current Iee(t) proportional to the absolute temperature T can be obtained at the collector of transistor Q13.
When using above current source, the conventional AGC detection circuit as shown in FIG. 1 is reduced the influence due to the temperature variation until the AGC output Vout becomes proportional to the absolute temperature T. However, the problem of the influence due to the temperature variation still remains. That is, the output current Iee(t) of the circuit shown in FIG. 2 is proportional to T, but not proportional to T.sup.2. So that the AGC output Vout is still suffered by the influence of the temperature variation.
As a method for further reducing the influence due to the temperature variation, it can be considered to operate second and third differential amplifiers D2 and D3 in FIG. 1 in switching operation. In this case, Equation (14) can be expressed as follows. ##EQU9##
In this manner, the influence due to the temperature can be reduced until AGC output Vout becomes proportional to absolute temperature T.
However, for switching differential circuits D2 and D3, it is necessary to produce a switching signal of a large amplitude completely synchronized with input signal Vin. For the purpose, for example, a phase locked loop (PLL) circuit is required, but resulting in complicating the circuit. For obtaining the switching signal comparatively easily, there is a method of obtaining the switching signal by amplifying input signal Vin in an additional circuit. In this method, however, the switching signal and input signal Vin are apt to shift in their phases with each other and an error is apt to arise in AGC output Vout. In this case, the error is especially increased in high frequency region. Further, for keeping the DC levels of the input signal and for switching signal in reasonable amplitudes, many level shift circuits are required, and resulting in suffered by temperature variations in these portions.
There is a further method for reducing the influence due to the temperature variation. In the method, resistors with high resistance sufficient to neglect thermal voltage Vt are connected to the emitters of respective transistors Q1 to Q6 in series.
Assuming the resistance of the resistors as Re, and Re&gt;&gt;2 Vt/Iee, AGC output Vout is expressed as follows. ##EQU10##
For satisfying Equation (22), the condition Re&gt;&gt;2. Vt/Iee is reguired as assuming in above. For example, when current Iee=1 mA, thermal voltage Vt=26 mV, and, 2.multidot.Vt/Iee.ltoreq.0.0 1Re (condition for neglecting thermal voltage Vt, where T=300 K.). In this case, the condition Re&gt;&gt;5.2 k is required. For obtaining AGC output Vout=0.1 V when DC level component of the input signal Vo=0.5 V, load resistance R1=86.5 k is required in the circuit of FIG. 1 when emitter resistances Re=5.2 k are used, while load resistance R1=4.32 k in the circuit of FIG. 1. In this case, assuming that current Iee=1 mA, the voltage drop in each resistor R1, R2 in the circuit of FIG. 1 is only 2.16 V. However, when adding emitter resistances R3=5.2 k to the emitters of each transistor Q1, Q2 in series, it reaches 47 V. Considering further the saturation of the transistors, necessary power supply voltage is only 8 V or so in the circuit in FIG. 1. However, when adding emitter resistances Re=5.2 k, power supply voltage V3 is required 55 V. Therefore, the method of reducing the influence due to temperature variation by adding emitter resistances Re is not practically useful.
As described above, in conventional AGC detection circuits, a sufficient countermeasure for preventing the influence due to the temperature variation can not be obtained and an AGC detection circuit of high-precision and usable over a wide range can not be obtained. Further, it is partially possible to take a countermeasure for preventing the influence due to the temperature variation. However, various problems such as complicity of circuits, increase of power consumption, and increase of power supply voltage are produced and resulting in reducing the range of use.